Field of the Invention
The present invention relates to a current measurement circuit.
Description of the Related Art
In order to analyze the base sequence of DNA (deoxyribonucleic acid), RNA (ribonucleic acid), or the like, a base sequence analyzing apparatus (sequencer) is employed. As a next-generation (fourth-generation) sequencer, various kinds of techniques have been sought by research institutions and industries. As one of such prospective techniques, the gating nanopore sequencing technique has attracted attention.
With the gating nanopore sequencing technique, DNA or RNA is moved such that it passes through a gap between a pair of nanometer-order electrodes (nano-electrodes). The tunnel current that flows through the electrode gap changes according to the base type (A, G, T, C) that passes through the electrode gap. The base sequence is determined based on the change in the tunnel current. This technique is anticipated to have the potential to provide a very low-cost and very compact-size apparatus that is capable of analyzing a base sequence. It should be noted that, in the present specification, examples of such a “nano-electrode” include sub-micro electrodes and micro electrodes having a larger size.
Also, as a method using a tunnel current in the same way as with the gating nanopore sequencing technique, the MCBJ (Mechanically Controllable Break Junction) method has been developed. With the MCBJ method, a nano-electrode is formed by breaking a metal wire.
As an important element technology, such a sequencer requires a current measurement device that is capable of measuring a tunnel current that flows through a nano-electrode gap with sufficiently high precision. That is to say, such a tunnel current has a current value on the order of several tens of pA. Accordingly, in order to judge the base type, there is a need to detect a difference in conductance on the order of several ps (picosecond).
The present inventors have investigated an arrangement employing a transimpedance amplifier as a microscopic current measurement device. FIG. 1 is a circuit diagram showing a current measurement circuit 900 including a transimpedance amplifier 800. The transimpedance amplifier 800 includes an operational amplifier 802 and a resistor RF arranged between the inverting input terminal (−) and the output terminal of the operational amplifier 802. A predetermined electric potential VREF (e.g., ground voltage) is input to the non-inverting input terminal (+) of the operational amplifier 802. A capacitor CF is connected in parallel with the resistor RF, in order to provide the circuit with stable operation.
A DUT (device under test) 810 includes a sample such as DNA, RNA, or the like (which will collectively be referred to as “DNA” hereafter), and a chip configured to house the sample. A nanochannel, nanopillar structure, and an electrode pair are formed in the chip such that a DNA molecule separated from the sample passes through such components. A cable 820 connects the DUT 810 and the transimpedance amplifier 800.
FIG. 2 is an equivalent circuit diagram showing an equivalent circuit of the current measurement circuit 900 shown in FIG. 1. The DUT 810 can be modeled as a circuit comprising a current source 812 that generates a tunnel current iDUT, a parasitic parallel resistor RDUT, and a parasitic parallel capacitor CDUT.
The cable 820 includes a first line 822 that connects one end 814 of the DUT 810 and the inverting input terminal of the operational amplifier 802, and a second line 824 that connects the other end 816 of the DUT 810 and the non-inverting input terminal of the operational amplifier 802. Here, CCAB represents a parasitic capacitance that occurs between the two lines 822 and 824. In a case in which the cable 820 is configured as a coaxial cable, a parasitic capacitance of 10 pF occurs in increments of 10 cm of the cable 820.
Various kinds of parasitic capacitances occur in the input stage of the transimpedance amplifier 800. For example, CPRO represents a parasitic capacitance that occurs due to an ESD protection element 830 such as a diode, ESD suppresser, or the like. The operational amplifier 802 is represented by an equivalent circuit comprising an ideal amplifier 804 and various kinds of parasitic capacitances. Here, CMN and CMP each represent a common input capacitance, and CD represents a differential input capacitance. It should be noted that, in FIG. 2, the resistance values and capacitance values are shown for exemplary purposes only.
The DC transimpedance of the transimpedance amplifier 800 is represented by the following Expression.20×log10(RF)(dB)  (1)
For example, in a case in which RF=1 GΩ, the transimpedance amplifier 800 has a DC transimpedance of 180 dB.
Such a DNA sequencer is required to identify the kinds of bases with respect to an enormous number of base pairs, the number of which is on the order of several billion. A fourth-generation DNA sequencer is required to provide a measurement time on the order of 1 ms per base. However, it is difficult for such a fourth-generation DNA sequencer to identify a base based on a single measurement due to the influence of noise. Thus, the tunnel current is measured multiple times during the measurement time of 1 ms, and the base is identified using a statistical method. Specifically, the base is identified based on a histogram of the measurement results, for example. For example, in a case in which the tunnel current is measured 100 times during the measurement time of 1 ms, such an arrangement requires a sampling rate of 100 ksps. In this case, the transimpedance amplifier is required to have a bandwidth of several hundreds of kHz to several MHz, which is estimated giving consideration to a margin.
Here, examination will be made regarding the frequency characteristics of the transimpedance amplifier 800 shown in FIG. 2. The cutoff frequency f2 is represented by the following Expression (2).f2=1/{2πRF×(CF+CS/AOL)}  (2)
It should be noted that CS is represented by CS=CDUT+CCAB+CPRO+CD. Here, AOL represents the open loop gain of the operational amplifier. As can be understood from Expression (2), in order to raise the cutoff frequency f2, an approach can be employed in which CF and CS are each reduced, and AOL is raised over a wide bandwidth. Here, CS will be referred to as “input shunt capacitance”. In a case in which the open loop gain AOL is sufficiently large, and the input shunt capacitance CS is sufficiently small, Expression (2) is approximated by the following Expression (3).f2≈1/{2πRF×CF}  (3)
For example, in a case in which RF=1 GΩ, and CF=10 fF, f2=15.9 kHz is obtained based on Expression (3).
However, the tunnel current has a very small current value. Thus, such a tunnel current is affected by measurement system noise, which is a problem. FIG. 3 is a diagram showing the noise characteristics of the transimpedance amplifier. In order to detect such a microscopic current, such an arrangement requires a resistor RF on the order of several tens of MΩ to several TΩ. Accordingly, in the low-frequency range, thermal noise due to the resistor RF becomes dominant.VNOISE=√(4×k×T×RF)
Here, T represents the temperature, and k represents the Boltzmann constant. This expression represents the voltage noise density per unit frequency.
In a case in which RF=1 GΩ, and T=27 degrees, VNOISE=4.1 μV/√Hz (which is also represented by “V/rtHz”) is obtained.
The transimpedance amplifier 800 imposes a band limit on the thermal noise with the aforementioned cutoff frequency f2 as the boundary. Thus, in the high-frequency range that is higher than the cutoff frequency f2, the noise from the transimpedance amplifier 800 becomes dominant as compared with the thermal noise that occurs in the resistor RF. In the high-frequency range, the noise gain of the amplifier is proportional to (CF+CS+CM+CD)/CF. Thus, in order to reduce the noise, CS, CM, and CD must be reduced, and CF must be raised. However, an increase in CF leads to a reduction in the cutoff frequency f2, which is opposite to a requirement of increasing the bandwidth. Thus, there is a need to design the capacitor CF to have as small a value as possible in a range so as to ensure system stability. As described above, with the transimpedance amplifier shown in FIG. 1, there is a tradeoff relation between the bandwidth (cutoff frequency) to be raised and the noise to be reduced. That is to say, it is difficult to provide both a wide bandwidth and low noise.
In particular, the operational amplifier has a very high input impedance. Accordingly, the transimpedance amplifier is greatly affected by electric-field noise. In order to reduce such noise, a technique is known in which the signal line is covered by a shield, and the electric potential at the shield is controlled. However, in a case in which such a shield is provided as an additional component to the transimpedance amplifier which is required to provide a high-speed operation, this leads to an increase in the parasitic capacitance due to the signal line. In addition, an amplifier that drives the shield involves a capacitance. Thus, such an arrangement leads to a narrow bandwidth and a reduced operation speed.